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AD8139


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Low Noise, Rail-to-Rail, Differential ADC Driver AD8139
FEATURES
Fully differential Low noise 2.25 nV/﹟Hz 2.1 pA/﹟Hz Low harmonic distortion 98 dBc SFDR @ 1 MHz 85 dBc SFDR @ 5 MHz 72 dBc SFDR @ 20 MHz High speed 410 MHz, 3 dB BW (G = 1) 800 V/米s slew rate 45 ns settling time to 0.01% 69 dB output balance @ 1 MHz 80 dB dc CMRR Low offset: ㊣0.5 mV maximum Low input offset current: 0.5 米A maximum Differential input and output Differential-to-differential or single-ended-to-differential operation Rail-to-rail output Adjustable output common-mode voltage Wide supply voltage range: 5 V to 12 V Available in a small SOIC package and an 8-lead LFCSP

APPLICATIONS
ADC drivers to 18 bits Single-ended-to-differential converters Differential filters Level shifters Differential PCB drivers Differential cable drivers

FUNCTIONAL BLOCK DIAGRAMS
每IN 1 VOCM 2 V+ 3 +OUT 4

AD8139
8 7 6 5

+IN NC V每
04679-001

每OUT

NC = NO CONNECT

Figure 1. 8-Lead SOIC

AD8139
TOP VIEW (Not to Scale) 每IN 1 VOCM 2 V+ 3 +OUT 4 NC = NO CONNECT 8 +IN 7 NC 6 V每 5 每OUT
04679-102

Figure 2. 8-Lead LFCSP

GENERAL DESCRIPTION
The AD8139 is an ultralow noise, high performance differential amplifier with rail-to-rail output. With its low noise, high SFDR, and wide bandwidth, it is an ideal choice for driving ADCs with resolutions to 18 bits. The AD8139 is easy to apply, and its internal common-mode feedback architecture allows its output common-mode voltage to be controlled by the voltage applied to one pin. The internal feedback loop also provides outstanding output balance as well as suppression of even-order harmonic distortion products. Fully differential and singleended-to-differential gain configurations are easily realized by the AD8139. Simple external feedback networks consisting of four resistors determine the closed-loop gain of the amplifier. The AD8139 is manufactured on the Analog Devices, Inc. proprietary, second-generation XFCB process, enabling it to achieve low levels of distortion with input voltage noise of only 2.25 nV/﹟Hz.

The AD8139 is available in an 8-lead SOIC package with an exposed paddle (EP) on the underside of its body and a 3 mm ℅ 3 mm LFCSP. It is rated to operate over the temperature range of ?40∼C to +125∼C.
100

INPUT VOLTAGE NOISE (nV/ Hz)

10

100

1k

10k 100k 1M FREQUENCY (Hz)

10M

100M

1G

Figure 3. Input Voltage Noise vs. Frequency

Rev. B
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ?2007 Analog Devices, Inc. All rights reserved.

04679-078

1 10

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AD8139 TABLE OF CONTENTS
Features .............................................................................................. 1 Applications....................................................................................... 1 Functional Block Diagrams............................................................. 1 General Description ......................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 VS = ㊣5 V, VOCM = 0 V .................................................................. 3 VS = 5 V, VOCM = 2.5 V ................................................................. 5 Absolute Maximum Ratings............................................................ 7 Thermal Resistance ...................................................................... 7 ESD Caution.................................................................................. 7 Pin Configurations and Function Descriptions ............................8 Typical Performance Characteristics ..............................................9 Test Circuits ................................................................................ 17 Theory of Operation ...................................................................... 18 Typical Connection and Definition of Terms ........................ 18 Applications..................................................................................... 19 Estimating Noise, Gain, and Bandwidth with Matched Feedback Networks .................................................................... 19 Outline Dimensions ....................................................................... 24 Ordering Guide .......................................................................... 24

REVISION HISTORY
10/07〞Rev. A to Rev. B. Changes to General Description .................................................... 1 Inserted Figure 2; Renumbered Sequentially................................ 1 Changes to Table 1............................................................................ 3 Changes to Table 2............................................................................ 5 Changes to Table 6 and Layout ....................................................... 8 Inserted Figure 6; Renumbered Sequentially................................ 8 Changes to Figure 30...................................................................... 12 Changes to Layout .......................................................................... 17 Changes to Figure 63...................................................................... 22 Changes to Exposed Paddle (EP) Section ................................... 23 Updated Outline Dimensions ....................................................... 24 8/04〞Rev. 0 to Rev. A. Added 8-Lead LFCSP.........................................................Universal Changes to General Description .....................................................1 Changes to Figure 2...........................................................................1 Changes to VS = ㊣5 V, VOCM = 0 V Specifications .........................3 Changes to VS = 5 V, VOCM = 2.5 V Specifications.........................5 Changes to Table 4.............................................................................7 Changes to Maximum Power Dissipation Section........................7 Changes to Figure 26 and Figure 29............................................. 12 Inserted Figure 39 and Figure 42.................................................. 14 Changes to Figure 45 to Figure 47................................................ 15 Inserted Figure 48........................................................................... 15 Changes to Figure 52 and Figure 53............................................. 16 Changes to Figure 55 and Figure 56............................................. 17 Changes to Table 6.......................................................................... 19 Changes to Voltage Gain Section ................................................. 19 Changes to Driving a Capacitive Load Section .......................... 22 Changes to Ordering Guide .......................................................... 24 Updated Outline Dimensions....................................................... 24 5/04〞Revision 0: Initial Version

Rev. B | Page 2 of 24

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AD8139 SPECIFICATIONS
VS = ㊣5 V, VOCM = 0 V
TA = 25∼C, differential gain = 1, RL, dm = 1 k次, RF = RG = 200 次, unless otherwise noted. TMIN to TMAX = ?40∼C to +125∼C. Table 1.
Parameter DIFFERENTIAL INPUT PERFORMANCE Dynamic Performance ?3 dB Small Signal Bandwidth ?3 dB Large Signal Bandwidth Bandwidth for 0.1 dB Flatness Slew Rate Settling Time to 0.01% Overdrive Recovery Time Noise/Harmonic Performance SFDR Conditions Min Typ Max Unit

VO, dm = 0.1 V p-p VO, dm = 2 V p-p VO, dm = 0.1 V p-p VO, dm = 2 V step VO, dm = 2 V step, CF = 2 pF G = 2, VIN, dm = 12 V p-p triangle wave VO, dm = 2 V p-p, fC = 1 MHz VO, dm = 2 V p-p, fC = 5 MHz VO, dm = 2 V p-p, fC = 20 MHz VO, dm = 2 V p-p, fC = 10.05 MHz ㊣ 0.05 MHz f = 100 kHz f = 100 kHz VIP = VIN = VOCM = 0 V TMIN to TMAX TMIN to TMAX

340 210

410 240 45 800 45 30 98 85 72 ?90 2.25 2.1

MHz MHz MHz V/米s ns ns dBc dBc dBc dBc nV/﹟Hz pA/﹟Hz +500 8.0 0.5 米V 米V/∼C 米A 米A dB V k次 M次 pF dB V V mA dB

Third-Order IMD Input Voltage Noise Input Current Noise DC Performance Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Offset Current Open-Loop Gain Input Characteristics Input Common-Mode Voltage Range Input Resistance Input Capacitance CMRR Output Characteristics Output Voltage Swing

?500

㊣150 1.25 2.25 0.12 114

?4 Differential Common mode Common mode ?VICM = ㊣1 V dc, RF = RG = 10 k次 Each single-ended output, RF = RG = 10 k次 Each single-ended output, RL, dm = open circuit, RF = RG = 10 k次 Each single-ended output f = 1 MHz 600 1.5 1.2 84

+4

80 ?VS + 0.20 ?VS + 0.15

+VS 每 0.20 +VS ? 0.15 100 ?69

Output Current Output Balance Error VOCM TO VO, cm PERFORMANCE VOCM Dynamic Performance ?3 dB Bandwidth Slew Rate Gain VOCM Input Characteristics Input Voltage Range Input Resistance Input Offset Voltage Input Voltage Noise Input Bias Current CMRR

VO, cm = 0.1 V p-p VO, cm = 2 V p-p 0.999 ?3.8 VOS, cm = VO, cm ? VOCM; VIP = VIN = VOCM = 0 V f = 100 kHz ?VOCM/?VO, dm, ?VOCM = ㊣1 V ?900

515 250 1.000

1.001 +3.8

MHz V/米s V/V V M次 米V nV/﹟Hz 米A dB

74

3.5 ㊣300 3.5 1.3 88

+900 4.5

Rev. B | Page 3 of 24

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AD8139
Parameter POWER SUPPLY Operating Range Quiescent Current +PSRR ?PSRR OPERATING TEMPERATURE RANGE Conditions Min +4.5 Change in +VS = ㊣1 V Change in ?VS = ㊣1 V 95 95 ?40 24.5 112 109 Typ Max ㊣6 25.5 Unit V mA dB dB ∼C

+125

Rev. B | Page 4 of 24

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AD8139
VS = 5 V, VOCM = 2.5 V
TA = 25∼C, differential gain = 1, RL, dm = 1 k次, RF = RG = 200 次, unless otherwise noted. TMIN to TMAX = ?40∼C to +125∼C. Table 2.
Parameter DIFFERENTIAL INPUT PERFORMANCE Dynamic Performance ?3 dB Small Signal Bandwidth ?3 dB Large Signal Bandwidth Bandwidth for 0.1 dB Flatness Slew Rate Settling Time to 0.01% Overdrive Recovery Time Noise/Harmonic Performance SFDR Conditions Min Typ Max Unit

VO, dm = 0.1 V p-p VO, dm = 2 V p-p VO, dm = 0.1 V p-p VO, dm = 2 V step VO, dm = 2 V step G = 2, VIN, dm = 7 V p-p triangle wave VO, dm = 2 V p-p, fC = 1 MHz VO, dm = 2 V p-p, fC = 5 MHz, RL = 800 次 VO, dm = 2 V p-p, fC = 20 MHz, RL = 800 次 VO, dm = 2 V p-p, fC = 10.05 MHz ㊣ 0.05 MHz f = 100 kHz f = 100 kHz VIP = VIN = VOCM = 2.5 V TMIN to TMAX TMIN to TMAX

330 135

385 165 34 540 55 35 99 87 75 ?87 2.25 2.1

MHz MHz MHz V/米s ns ns dBc dBc dBc dBc nV/﹟Hz pA/﹟Hz +500 7.5 0.5 米V 米V/∼C 米A 米A dB V k次 M次 pF dB V V mA dB

Third-Order IMD Input Voltage Noise Input Current Noise DC Performance Input Offset Voltage Input Offset Voltage Drift Input Bias Current Input Offset Current Open-Loop Gain Input Characteristics Input Common-Mode Voltage Range Input Resistance Input Capacitance CMRR Output Characteristics Output Voltage Swing

?500

㊣150 1.25 2.2 0.13 112

1 Differential Common mode Common mode 忖VICM = ㊣1 V dc, RF = RG = 10 k次 Each single-ended output, RF = RG = 10 k次 Each single-ended output, RL, dm = open circuit, RF = RG = 10 k次 Each single-ended output f = 1 MHz 600 1.5 1.2 79

4

75 ?VS + 0.15 ?VS + 0.10

+VS ? 0.15 +VS ? 0.10 80 ?70

Output Current Output Balance Error VOCM TO VO, cm PERFORMANCE VOCM Dynamic Performance ?3 dB Bandwidth Slew Rate Gain VOCM Input Characteristics Input Voltage Range Input Resistance Input Offset Voltage Input Voltage Noise Input Bias Current CMRR

VO, cm = 0.1 V p-p VO, cm = 2 V p-p 0.999 1.0 VOS, cm = VO, cm ? VOCM; VIP = VIN = VOCM = 2.5 V f = 100 kHz 忖VOCM/忖VO, dm, 忖VOCM = ㊣1 V ?1.0

440 150 1.000

1.001 3.8

MHz V/米s V/V V M次 mV nV/﹟Hz 米A dB

67

3.5 ㊣0.45 3.5 1.3 79

+1.0 4.2

Rev. B | Page 5 of 24

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AD8139
Parameter POWER SUPPLY Operating Range Quiescent Current +PSRR ?PSRR OPERATING TEMPERATURE RANGE Conditions Min +4.5 Change in +VS = ㊣1 V Change in ?VS = ㊣1 V 86 92 ?40 21.5 97 105 Typ Max ㊣6 22.5 Unit V mA dB dB ∼C

+125

Rev. B | Page 6 of 24

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AD8139 ABSOLUTE MAXIMUM RATINGS
Table 3.
Parameter Supply Voltage VOCM Power Dissipation Input Common-Mode Voltage Storage Temperature Range Operating Temperature Range Lead Temperature (Soldering 10 sec) Junction Temperature Rating 12 V ㊣VS See Figure 4 ㊣VS ?65∼C to +125∼C ?40∼C to +125∼C 300∼C 150∼C

The power dissipated in the package (PD) is the sum of the quiescent power dissipation and the power dissipated in the package due to the load drive for all outputs. The quiescent power is the voltage between the supply pins (VS) times the quiescent current (IS). The load current consists of differential and common-mode currents flowing to the load, as well as currents flowing through the external feedback networks and the internal common-mode feedback loop. The internal resistor tap used in the common-mode feedback loop places a 1 k次 differential load on the output. RMS output voltages should be considered when dealing with ac signals. Airflow reduces 牟JA. In addition, more metal directly in contact with the package leads from metal traces, through holes, ground, and power planes reduce the 牟JA. Figure 4 shows the maximum safe power dissipation in the package vs. the ambient temperature for the exposed paddle (EP) 8-lead SOIC (牟JA = 70∼C/W) and the 8-lead LFCSP (牟JA = 70∼C/W) on a JEDEC standard 4-layer board. 牟JA values are approximations.
4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5
04679-055

Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

THERMAL RESISTANCE
牟JA is specified for the worst-case conditions, that is, 牟JA is specified for device soldered in circuit board for surface-mount packages. Table 4.
Package Type 8-Lead SOIC with EP/4-Layer 8-Lead LFCSP/4-Layer 牟JA 70 70 Unit ∼C/W ∼C/W

MAXIMUM POWER DISSIPATION (W)

Maximum Power Dissipation
The maximum safe power dissipation in the AD8139 package is limited by the associated rise in junction temperature (TJ) on the die. At approximately 150∼C, which is the glass transition temperature, the plastic will change its properties. Even temporarily exceeding this temperature limit can change the stresses that the package exerts on the die, permanently shifting the parametric performance of the AD8139. Exceeding a junction temperature of 175∼C for an extended period can result in changes in the silicon devices potentially causing failure.

SOIC AND LFCSP

0 每40

每20

0

20

40

60

80

100

120

AMBIENT TEMPERATURE (∼C)

Figure 4. Maximum Power Dissipation vs. Temperature for a 4-Layer Board

ESD CAUTION

Rev. B | Page 7 of 24

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AD8139 PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
AD8139
每IN 1 VOCM 2 V+ 3 +OUT 4

AD8139
8 7 6 5

TOP VIEW (Not to Scale)
+IN NC V每
04679-003

每IN 1 VOCM 2 V+ 3 +OUT 4 NC = NO CONNECT

8 +IN 7 NC 6 V每 5 每OUT
04679-103

每OUT

NC = NO CONNECT

Figure 5. 8-Lead SOIC Pin Configuration

Figure 6. 8-Lead LFCSP Pin Configuration

Table 5. Pin Function Descriptions
Pin No. 1 2 3 4 5 6 7 8 9 Mnemonic ?IN VOCM V+ +OUT ?OUT V? NC +IN Exposed Paddle Description Inverting Input. An internal feedback loop drives the output common-mode voltage to be equal to the voltage applied to the VOCM pin, provided the operation of the amplifier remains linear. Positive Power Supply Voltage. Positive Side of the Differential Output. Negative Side of the Differential Output. Negative Power Supply Voltage. No Internal Connection. Noninverting Input. Solder exposed paddle on back of package to ground plane or to a power plane.

Rev. B | Page 8 of 24

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AD8139 TYPICAL PERFORMANCE CHARACTERISTICS
Unless otherwise noted, differential gain = +1, RG = RF = 200 次, RL, dm = 1 k次, VS = ㊣5 V, TA = 25∼C, VOCM = 0 V. Refer to the basic test circuit in Figure 57 for the definition of terms.
2 1 2 G=1 G=2 1 G=1

NORMALIZED CLOSED-LOOP GAIN (dB)

0 每1 每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 每13 1 RG = 200? VO, dm = 0.1V p-p G = 10

NORMALIZED CLOSED-LOOP GAIN (dB)

0 每1 每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 每13 1 RG = 200? VO, dm = 2.0V p-p 10 100 1000
04679-007 04679-009 04679-008

G=5

G=2

G=5

G = 10

10

100

1000

FREQUENCY (MHz)

04679-004

FREQUENCY (MHz)

Figure 7. Small Signal Frequency Response for Various Gains
5 4 3 2 VS = +5V

Figure 10. Large Signal Frequency Response for Various Gains
3 2 1 0

CLOSED-LOOP GAIN (dB)

0 每1 每2 每3 每4 每5 每6 每7 每8 每9 每10 10 VO, dm = 0.1V p-p 100

CLOSED-LOOP GAIN (dB)

1 VS = ㊣5V

每1 每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 VO, dm = 2.0V p-p 100 FREQUENCY (MHz) 1000 每12 10 VS = +5V VS = ㊣5V

1000

FREQUENCY (MHz)

Figure 8. Small Signal Frequency Response for Various Power Supplies
3 2 1 0

04679-005

Figure 11. Large Signal Frequency Response for Various Power Supplies
3 2 1 0

+125∼C +85∼C

+125∼C

+85∼C

CLOSED-LOOP GAIN (dB)

每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 10 VO, dm = 0.1V p-p 100 FREQUENCY (MHz) +25∼C 1000
04679-006

CLOSED-LOOP GAIN (dB)

每1

每1 每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 10 VO, dm = 2.0V p-p 100 FREQUENCY (MHz) 1000 每40∼C +25∼C

每40∼C

Figure 9. Small Signal Frequency Response at Various Temperatures

Figure 12. Large Signal Frequency Response at Various Temperatures

Rev. B | Page 9 of 24

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AD8139
3 2 1 0 RL = 200? RL = 100?

2 1 0 每1
CLOSED-LOOP GAIN (dB)

RL = 100? RL = 500?

CLOSED-LOOP GAIN (dB)

每1 每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 10 VO, dm = 0.1V p-p 100 FREQUENCY (MHz) RL = 1k?
04679-040

每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 VO, dm = 2.0V p-p 每13 10 RL = 200? 100 FREQUENCY (MHz) 1000
04679-041
04679-0-042

RL = 500?

RL = 1k?

1000

Figure 13. Small Signal Frequency Response for Various Loads
3 2 1 0

Figure 16. Large Signal Frequency Response for Various Loads
2 1 0 每1

CF = 0pF CF = 1pF

CF = 0pF

CF = 1pF

CLOSED-LOOP GAIN (dB)

CLOSED-LOOP GAIN (dB)

每1 每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 10 VO, dm = 0.1V p-p
04679-011

每2 每3 每4 每5 每6 每7 每8 每9 每10 每11 每12 VO, dm = 2.0V p-p 每13 10

CF = 2pF

CF = 2pF

100 FREQUENCY (MHz)

1000

100 FREQUENCY (MHz)

1000

Figure 14. Small Signal Frequency Response for Various CF
6 5 4 3 VOCM = +4.3V VOCM = 每4.3V VOCM = 每4V VOCM = +4V
NORMALIZED CLOSED-LOOP GAIN (dB)

Figure 17. Large Signal Frequency Response for Various CF
0.5 0.4 0.3 0.2 0.1 0 每0.1 每0.2 每0.3 每0.4 每0.5 1 10 FREQUENCY (Hz) RL = 1k? (VO, dm = 0.1V p-p) RL = 100? (VO, dm = 0.1V p-p) RL = 100? (VO, dm = 2.0V p-p) RL = 1k? (VO, dm = 2.0V p-p)

CLOSED-LOOP GAIN (dB)

2 1 0 每1 每2 每3 每4 每5 每6 每7 每8 每9 10 VO, dm = 0.1V p-p VOCM = 0V

04679-012

100 FREQUENCY (MHz)

1000

100

Figure 15. Small Signal Frequency Response at Various VOCM

Figure 18. 0.1 dB Flatness for Various Loads and Output Amplitudes

Rev. B | Page 10 of 24

04679-014

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AD8139
每30 每40 每50 VO, dm = 2.0V p-p 每30 每40 每50
DISTORTION (dBc)

VO, dm = 2.0V p-p

VS = +5V

DISTORTION (dBc)

每60 每70 每80 每90 每100 每110 每120

VS = ㊣5V

每60 每70 每80 每90 每100 每110 每120 VS = ㊣5V

VS = +5V

04679-015

1

10 FREQUENCY (MHz)

100

1

10 FREQUENCY (MHz)

100

Figure 19. Second Harmonic Distortion vs. Frequency and Supply Voltage
每30 每40 每50 每60
DISTORTION (dB)

Figure 22. Third Harmonic Distortion vs. Frequency and Supply Voltage
每30

VO, dm = 2.0V p-p

每40 每50 G=1 每60
DISTORTION (dB)

VO, dm = 2.0V p-p

每70 每80 每90 每100 每110 每120 每130
04679-016

每70 每80 每90 每100 每110 每120 每130 G=5 1 10 FREQUENCY (MHz) 100
04679-019 04679-020

G=5

G=2

G=1

G=2

每140 0.1

1

10 FREQUENCY (MHz)

100

每140 0.1

Figure 20. Second Harmonic Distortion vs. Frequency and Gain
每30 每40 每50
DISTORTION (dBc) DISTORTION (dBc)

Figure 23. Third Harmonic Distortion vs. Frequency and Gain
每30 每40 每50 RL = 100? RL = 200?

VO, dm = 2.0V p-p

VO, dm = 2.0V p-p

每60 每70 每80 每90 每100 每110 每120 1

RL = 100? RL = 200?

每60 每70 每80 每90 每100 每110 每120

RL = 500? RL = 1k?

RL = 500?

RL = 1k? 1 10 FREQUENCY (MHz) 100

10 FREQUENCY (MHz)

100

Figure 21. Second Harmonic Distortion vs. Frequency and Load

04679-017

每130 0.1

每130 0.1

Figure 24. Third Harmonic Distortion vs. Frequency and Load

Rev. B | Page 11 of 24

04679-018

每130 0.1

每130 0.1

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AD8139
每30 每40 每50
DISTORTION (dBc) DISTORTION (dBc)
每30

VO, dm = 2.0V p-p

每40 每50 每60 每70 每80 每90 每100 每110

VO, dm = 2.0V p-p

每60 每70 每80 每90 每100 每110 每120 每130 0.1 1 RF = 1k? RF = 200? RF = 500?

RF = 200?

RF = 1k? RF = 500? 1 10 FREQUENCY (MHz) 100
04679-024 04679-026

每120

10 FREQUENCY (MHz)

100

Figure 25. Second Harmonic Distortion vs. Frequency and RF
每80 FC = 2MHz 每90 VS = +5V 每100 VS = ㊣5V

04679-021

每130 0.1

Figure 28. Third Harmonic Distortion vs. Frequency and RF
每80 每90 VS = +5V 每100 FC = 2MHz

DISTORTION (dBc)

DISTORTION (dBc)

VS = ㊣5V 每110 每120 每130 每140 每150

每110 每120 每130 每140 每150

04679-022

0

1

2

3

4 VO, dm (V p-p)

5

6

7

8

0

1

2

3

4 VO, dm (V p-p)

5

6

7

8

Figure 26. Second Harmonic Distortion vs. Output Amplitude
每60 每70 每80

Figure 29. Third Harmonic Distortion vs. Output Amplitude
每60 每70 每80
DISTORTION (dBc)

VO, dm = 2V p-p FC = 2MHz

VO, dm = 2V p-p FC = 2MHz

DISTORTION (dBc)

每90 每100 每110 每120

SECOND HARMONIC

每90 每100 每110 每120

SECOND HARMONIC

THIRD HARMONIC 0 0.5 1.0 1.5 2.0 2.5 VOCM (V) 3.0 3.5 4.0 4.5 5.0
04679-023

THIRD HARMONIC 每130 每5 每4 每3 每2 每1 0 VOCM (V) 1 2 3 4 5

每130

Figure 27. Harmonic Distortion vs. VOCM, VS = +5 V

Figure 30. Harmonic Distortion vs. VOCM, VS = ㊣5 V

Rev. B | Page 12 of 24

04679-025

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AD8139
100 75 50 25 VO, dm = 100mV p-p 2.5 2.0 1.5 CF = 0pF (CF = 0pF, VS = ㊣5V) VO, dm (CF = 2pF, VS = ㊣5V) 1.0 CF = 2pF CF = 0pF CF = 2pF 2V p-p CF = 0pF 4V p-p

VO, dm (V)

0 每25 每50

VO, dm (V)
5ns/DIV
04679-043

0.5 0

每0.5 每1.0 每1.5

每75 每100 TIME (ns)

每2.0 每2.5 TIME (ns)
04679-044

5ns/DIV

Figure 31. Small Signal Transient Response for Various CF
0.100 0.075 0.050 0.025 RS = 31.6? CL, dm = 30pF

Figure 34. Large Signal Transient Response for Various CF
1.5 RS = 63.4? CL, dm = 15pF

1.0

0.5

VO, dm (V)

VO, dm (V)

RS = 63.4? CL, dm = 15pF

RS = 31.6? CL, dm = 30pF

0 每0.025 每0.050 每0.075 5ns/DIV
04679-064

0

每0.5

每1.0 5ns/DIV 每1.5 TIME (ns)
04679-065

每0.100 TIME (ns)

Figure 32. Small Signal Transient Response for Capacitive Loads
5 0 VO, dm = 2V p-p 每5 FC1 = 10MHz 每10 FC2 = 10.1MHz 每15 每20 每25 每30 每35 每40 每45 每50 每55 每60 每65 每70 每75 每80 每85 每90 每95 每100 9.55 9.65 9.75 9.85

Figure 35. Large Signal Transient Response for Capacitive Loads
1.5 600

CF = 2pF VO, dm = 2.0V p-p

1.0

400 ERROR (?V) 1DIV = 0.01%
04679-034

NORMALIZED OUTPUT (dBc)

AMPLITUDE (V)

0.5

200

0 ERROR 每0.5 VO, dm

0

每200

每1.0 VIN TIME (ns) 35ns/DIV

每400

9.95 10.05 10.15 10.25 10.35 10.45 10.55 FREQUENCY (MHz)

04679-027

每1.5

每600

Figure 33. Intermodulation Distortion

Figure 36. Settling Time (0.01%)

Rev. B | Page 13 of 24

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AD8139
1.5 ㊣5V

6 5 4 3
+5V

1.0

VS = +5V

0.5

CLOSED-LOOP GAIN (dB)

2 1 0 每1 每2 每3 每4 每5 每6 每7 VO, cm = 2.0V p-p VO, cm = 0.1V p-p VS = ㊣5V VS = ㊣5V

VOCM (V)

0

每0.5 VO, cm = 2V p-p VIN, dm = 0V 10ns/DIV 每1.5 TIME (ns)
04679-069

每1.0

每8 每9 10

VS = +5V 100 FREQUENCY (MHz) 1000
04679-038 04679-080 04679-045

Figure 37. VOCM Large Signal Transient Response
0 每10 每20 0 每10 每20

Figure 40. VOCM Frequency Response for Various Supplies

VIN, cm = 0.2V p-p INPUT CMRR = 忖VO, cm/忖VIN, cm

VO, cm = 0.2V p-p VOCM CMRR = 忖VO, dm/忖VO, cm

VOCM CMRR (dB)
04679-066

每30

每30 每40 每50 每60 每70 每80 每90

CMRR (dB)

每40 每50 每60 每70 每80 每90

RF = RG = 10k?

RF = RG = 200?

1

10 FREQUENCY (MHz)

100

500

1

10 FREQUENCY (MHz)

100

500

Figure 38. CMRR vs. Frequency
100 100

Figure 41. VOCM CMRR vs. Frequency

INPUT VOLTAGE NOISE (nV/ Hz)

VOCM VOLTAGE NOISE (nV/ Hz)
04679-079

10

10

1 10

100

1k

10k 100k 1M FREQUENCY (Hz)

10M

100M

1G

1 10

100

1k

10k 100k 1M FREQUENCY (Hz)

10M

100M

1G

Figure 39. Input Voltage Noise vs. Frequency

Figure 42. VOCM Voltage Noise vs. Frequency

Rev. B | Page 14 of 24

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AD8139
RL, dm = 1k? 每10 PSRR = 忖VO, dm/忖VS 每20 每30 0
14 12 10 8 6 4 2 0 每2 每4 每6 每8 每10 每12 G=2 2 ℅ VIN, dm VO, dm

PSRR (dB)

每40 每PSRR 每50 +PSRR 每60 每70 每80 每90 1 10 FREQUENCY (MHz) 100 500
04679-047

VOLTAGE (V)

每100

每14 TIME (ns)

Figure 43. PSRR vs. Frequency
100 VS = +5V 0 每10 每20 每30 每40 每50 每60 每70 0.01 0.1 每80

Figure 46. Overdrive Recovery

VO, dm = 1V p-p OUTPUT BALANCE = 忖VO, cm/忖VO, dm

OUTPUT IMPEDANCE (?)

VS = ㊣5V 1

0.1

OUTPUT BALANCE (dB)

10

04679-028

1

10 FREQUENCY (MHz)

100

1000

1

10 FREQUENCY (MHz)

100

500

Figure 44. Single-Ended Output Impedance vs. Frequency
700 600 500 300 200 100 0 每100 每200 每300 每400 每500 每600
04679-068

Figure 47. Output Balance vs. Frequency
300 VS = ㊣5V G = 1 (RF = RG = 200?) RL, dm = 1k? VS+ 每 VOP 每50

SINGLE-ENDED OUTPUT SWING FROM RAIL (mV)

VS+ 每 VOP

200

每150

VS = ㊣5V

VS = +5V

150

每200

VON 每 VS每

100

VON 每 VS每

每250

1k RESISTIVE LOAD (?)

10k

每20

0

20

40

60

80

100

120

TEMPERATURE (∼C)

Figure 45. Output Saturation Voltage vs. Output Load

Figure 48. Output Saturation Voltage vs. Temperature

Rev. B | Page 15 of 24

04679-077

每700 100

50 每40

每300

VON SWING FROM RAIL (mV)

VOP SWING FROM RAIL (mV)

400

250

每100

04679-067

04679-046

50ns/DIV

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AD8139
3.0 IOS
INPUT BIAS CURRENT (?A)
SUPPLY CURRENT (mA)

170

26 VS = ㊣5V 25
OFFSET CURRENT (nA)

2.5 IBIAS 2.0

145

24

120

23 VS = +5V 22

1.5

95

21

每20

0

20

40

60

80

100

120

每20

0

20

40

60

80

100

120

TEMPERATURE (∼C)

TEMPERATURE (∼C)

Figure 49. Input Bias and Offset Current vs. Temperature
10 8 6
INPUT BIAS CURRENT (?A)

Figure 52. Supply Current vs. Temperature
300 VOS, cm 250 400 600

4 2 0 每2 每4 每6 每8 每4

VS = ㊣5V VS = +5V
VOS, dm (?V)

200

200
VOS, cm (?V)
04679-071 04679-061

150 VOS, dm 100

0

每200

50

每400

每3

每2

每1

0 VACM (V)

1

2

3

4

5

04679-073

每10 每5

0 每40

每20

0

20

40

60

80

100

120

每600

TEMPERATURE (∼C)

Figure 50. Input Bias Current vs. Input Common-Mode Voltage
5 4 3 2 VS = ㊣2.5V

Figure 53. Offset Voltage vs. Temperature
50 45 40 35 FREQUENCY
VS = ㊣5V

COUNT = 350 MEAN = 每50?V STD DEV = 100?V

VOUT, cm (V)

1 0 每1 每2 每3 每4 每4 每3 每2 每1 0 VOCM (V) 1 2

30 25 20 15 10 5

3

4

5

04679-048

Figure 51. VOUT, cm vs. VOCM Input Voltage

Rev. B | Page 16 of 24

每500 每450 每400 每350 每300 每250 每200 每150 每100 每50 0 50 100 150 200 250 300 350 400 450 500 VOS, dm (?V)

每5 每5

0

Figure 54. VOS, dm Distribution

04679-060

04679-062

1.0 每40

70

20 每40

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AD8139
1.7 1.6 1.5
VOCM BIAS CURRENT (?A) VOCM BIAS CURRENT (?A)

6

4 VS = ㊣5V 2

1.4 1.3 1.2 1.1 1.0 0.9 0.8
04679-063

VS = +5V

0

每2

每4

每20

0

20

40

60

80

100

120

每4

每3

每2

每1

0 VOCM (V)

1

2

3

4

5

TEMPERATURE (∼C)

Figure 55. VOCM Bias Current vs. Temperature

Figure 56. VOCM Bias Current vs. VOCM Input Voltage

TEST CIRCUITS
RF 50? VTEST TEST SIGNAL SOURCE 60.4? 60.4? 50? VOCM RG = 200? RG = 200? CF AD8139 CF
04679-072

RL, dm = 1k?

每 VO, dm +

RF

Figure 57. Basic Test Circuit

50? VTEST TEST SIGNAL SOURCE 60.4? 60.4? 50? VOCM RG = 200? RG = 200?

RF = 200? RS AD8139 RS RF = 200? CL, dm 每 RL, dm VO, dm +
04679-075

Figure 58. Capacitive Load Test Circuit, G = +1

Rev. B | Page 17 of 24

04679-074

0.7 每40

每6 每5

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AD8139 THEORY OF OPERATION
The AD8139 is a high speed, low noise differential amplifier fabricated on the Analog Devices second-generation eXtra Fast Complementary Bipolar (XFCB) process. It is designed to provide two closely balanced differential outputs in response to either differential or single-ended input signals. Differential gain is set by external resistors, similar to traditional voltagefeedback operational amplifiers. The common-mode level of the output voltage is set by a voltage at the VOCM pin and is independent of the input common-mode voltage. The AD8139 has an H-bridge input stage for high slew rate, low noise, and low distortion operation and rail-to-rail output stages that provide maximum dynamic output range. This set of features allows for convenient single-ended-to-differential conversion, a common need to take advantage of modern high resolution ADCs with differential inputs.

outputs of identical amplitude and exactly 180∼ out of phase. The output balance performance does not require tightly matched external components, nor does it require that the feedback factors of each loop be equal to each other. Low frequency output balance is limited ultimately by the mismatch of an on-chip voltage divider, which is trimmed for optimum performance. Output balance is measured by placing a well-matched resistor divider across the differential voltage outputs and comparing the signal at the midpoint of the divider with the magnitude of the differential output. By this definition, output balance is equal to the magnitude of the change in output common-mode voltage divided by the magnitude of the change in output differential-mode voltage:
Output Balance = 忖VO, cm 忖VO, dm (3)

TYPICAL CONNECTION AND DEFINITION OF TERMS
Figure 59 shows a typical connection for the AD8139, using matched external RF/RG networks. The differential input terminals of the AD8139, VAP and VAN, are used as summing junctions. An external reference voltage applied to the VOCM terminal sets the output common-mode voltage. The two output terminals, VOP and VON, move in opposite directions in a balanced fashion in response to an input signal.
CF

The block diagram of the AD8139 in Figure 60 shows the external differential feedback loop (RF/RG networks and the differential input transconductance amplifier, GDIFF) and the internal common-mode feedback loop (voltage divider across VOP and VON and the common-mode input transconductance amplifier, GCM). The differential negative feedback drives the voltages at the summing junctions VAN and VAP to be essentially equal to each other. VAN = VAP (4) The common-mode feedback loop drives the output commonmode voltage, sampled at the midpoint of the two 500 次 resistors, to equal the voltage set at the VOCM terminal. This ensures that
VOP = VOCM + and VO, dm 2
VO, dm 2
RF 10pF

RF VIP VOCM VIN RG VAN RG VAP + VON 每 RL, dm VO, dm VOP RF
04679-050

AD8139


+

(5)

CF

VON = VOCM ?
VIN RG

(6)

Figure 59. Typical Connection

The differential output voltage is defined as VO, dm = VOP ? VON Common-mode voltage is the average of two voltages. The output common-mode voltage is defined as (1)

+

GO 500? MIDSUPPLY

VOP

VO, cm =

VOP + VON 2

VAN

(2)

VAP

GDIFF

GCM

500? VOCM VON

Output Balance
Output balance is a measure of how well VOP and VON are matched in amplitude and how precisely they are 180∼ out of phase with each other. It is the internal common-mode feedback loop that forces the signal component of the output common-mode towards zero, resulting in the near perfectly balanced differential
VIP

+

GO

RG

RF

Figure 60. Block Diagram

Rev. B | Page 18 of 24

04679-051

10pF

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AD8139 APPLICATIONS
ESTIMATING NOISE, GAIN, AND BANDWIDTH WITH MATCHED FEEDBACK NETWORKS
Estimating Output Noise Voltage
The total output noise is calculated as the root-sum-squared total of several statistically independent sources. Because the sources are statistically independent, the contributions of each must be individually included in the root-sum-square calculation. Table 6 lists recommended resistor values and estimates of bandwidth and output differential voltage noise for various closed-loop gains. For most applications, 1% resistors are sufficient.
Table 6. Recommended Values of Gain-Setting Resistors and Voltage Noise for Various Closed-Loop Gains
Gain 1 2 5 10 RG (次) 200 200 200 200 RF (次) 200 400 1k 2k 3 dB Bandwidth (MHz) 400 160 53 26 Total Output Noise (nV/﹟Hz) 5.8 9.3 19.7 37

Voltage Gain
The behavior of the node voltages of the single-ended-todifferential output topology can be deduced from the previous definitions. Referring to Figure 59, (CF = 0) and setting VIN = 0, one can write
VIP ? VAP VAP ? VON = RG RF

(11) (12)

? RG ? VAN = VAP = VOP ? ? ? RF + RG ?

Solving the above two equations and setting VIP to Vi gives the gain relationship for VO, dm/Vi.
VOP ? VON = VO, dm = RF V RG i

(13)

An inverting configuration with the same gain magnitude can be implemented by simply applying the input signal to VIN and setting VIP = 0. For a balanced differential input, the gain from VIN, dm to VO, dm is also equal to RF/RG, where VIN, dm = VIP ? VIN.

Feedback Factor Notation
When working with differential amplifiers, it is convenient to introduce the feedback factor 汕, which is defined as

The differential output voltage noise contains contributions from the input voltage noise and input current noise of the AD8139 as well as those from the external feedback networks. The contribution from the input voltage noise spectral density is computed as

汕=

RG RF + RG

(14)

? R ? Vo_n1 = vn ?1 + F ? , or equivalently, vn/汕 ? R ? G ? ?

(7)

This notation is consistent with conventional feedback analysis and is very useful, particularly when the two feedback loops are not matched.

Input Common-Mode Voltage
The linear range of the VAN and VAP terminals extends to within approximately 1 V of either supply rail. Because VAN and VAP are essentially equal to each other, they are both equal to the input common-mode voltage of the amplifier. Their range is indicated in the Specifications tables as input common-mode range. The voltage at VAN and VAP for the connection diagram in Figure 59 can be expressed as VAN = VAP = VACM =
(V + VIN ) ? ? RG ? ? RF ℅ VOCM ? ℅ IP ? ?+? RF + RG 2 RF + RG ? ? ? ? (15)

where vn is defined as the input-referred differential voltage noise. This equation is the same as that of traditional op amps. The contribution from the input current noise of each input is computed as Vo_n2 = in (RF) where in is defined as the input noise current of one input. Each input needs to be treated separately because the two input currents are statistically independent processes. The contribution from each RG is computed as (8)

?R Vo_n3 = 4kTRG ? F ?R ? G

? ? ? ?

(9)

where VACM is the common-mode voltage present at the amplifier input terminals. Using the 汕 notation, Equation 15 can be written as follows: VACM = 汕VOCM + (1 ? 汕)VICM (16) (17) or equivalently, VACM = VICM + 汕(VOCM ? VICM) where VICM is the common-mode voltage of the input signal, that is, VICM = VIP + VIN/2.

This result can be intuitively viewed as the thermal noise of each RG multiplied by the magnitude of the differential gain. The contribution from each RF is computed as Vo_n4 = ﹟4kTRF (10)

Rev. B | Page 19 of 24

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AD8139
For proper operation, the voltages at VAN and VAP must stay within their respective linear ranges.

Calculating Input Impedance
The input impedance of the circuit in Figure 59 depends on whether the amplifier is being driven by a single-ended or a differential signal source. For balanced differential input signals, the differential input impedance (RIN, dm) is simply RIN, dm = 2RG (18) For a single-ended signal (for example, when VIN is grounded and the input signal drives VIP), the input impedance becomes

The input impedance of a conventional inverting op amp configuration is simply RG, but it is higher in Equation 19 because a fraction of the differential output voltage appears at the summing junctions, VAN and VAP. This voltage partially bootstraps the voltage across the input resistor RG, leading to the increased input resistance.

Input Common-Mode Swing Considerations
In some single-ended-to-differential applications, when using a single-supply voltage, attention must be paid to the swing of the input common-mode voltage, VACM. Consider the case in Figure 61, where VIN is 5 V p-p swinging about a baseline at ground, and VREF is connected to ground. The circuit has a differential gain of 1.6 and 汕 = 0.38. VICM has an amplitude of 2.5 V p-p and is swinging about ground. Using the results in Equation 16, the common-mode voltage at the inputs of the AD8139, VACM, is a 1.5 V p-p signal swinging about a baseline of 0.95 V. The maximum negative excursion of VACM in this case is 0.2 V, which exceeds the lower input common-mode voltage limit.
5V

RIN =

RG RF 1? 2(RG + RF )

(19)

0.1?F 324? 200? 2.5V VIN VREF VOCM 3 8 2 1 + 5 15? 2.7nF

0.1?F

20?

0.1?F

AVDD IN每

DVDD

+2.5V GND 每2.5V

AD8139
每 6 4 324? +1.7V +0.95V +0.2V 15?

AD7674
IN+ DGND AGND REFGND REF REFBUFIN PDBUF 47?F

200?

2.7nF

VACM WITH VREF = 0

0.1?F

Figure 61. AD8139 Driving AD7674, 18-Bit, 800 kSPS ADC

Rev. B | Page 20 of 24

04679-052

ADR431 2.5V REFERENCE

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AD8139
One way to avoid the input common-mode swing limitation is to bias VIN and VREF at midsupply. In this case, VIN is 5 V p-p swinging about a baseline at 2.5 V, and VREF is connected to a low-Z 2.5 V source. VICM now has an amplitude of 2.5 V p-p and is swinging about 2.5 V. Using the results in Equation 17, VACM is calculated to be equal to VICM because VOCM = VICM. Therefore, VACM swings from 1.25 V to 3.75 V, which is well within the input common-mode voltage limits of the AD8139. Another benefit seen in this example is that because VOCM = VACM = VICM no wasted common-mode current flows. Figure 62 illustrates how to provide the low-Z bias voltage. For situations that do not require a precise reference, a simple voltage divider suffices to develop the input voltage to the buffer.
5V 0.1?F 200? VIN 0V TO 5V VOCM 3 8 2 1 + 5

Estimating DC Errors
Primary differential output offset errors in the AD8139 are due to three major components: the input offset voltage, the offset between the VAN and VAP input currents interacting with the feedback network resistances, and the offset produced by the dc voltage difference between the input and output common-mode voltages in conjunction with matching errors in the feedback network. The first output error component is calculated as ? R + RG Vo _ e1 = VIO ? F ? R G ? ? ? , or equivalently as VIO/汕 ? ? (21)

where VIO is the input offset voltage. The input offset voltage of the AD8139 is laser trimmed and guaranteed to be less than 500 米V.
324?

The second error is calculated as
? R + RG Vo _ e2 = I IO ? F ? R G ? ?? RG RF ?? ?? R + R G ?? F ? ? = I IO (RF ) ? ?
(22)

AD8139
每 6 4 324? 5V TO AD7674 REFBUFIN

200? 0.1?F 0.1?F

where IIO is defined as the offset between the two input bias currents. The third error voltage is calculated as Vo_e3 = 忖enr ℅ (VICM ? VOCM) (23) where 忖enr is the fractional mismatch between the two feedback resistors. The total differential offset error is the sum of these three error sources.
04679-053

10?F

+

+

AD8031


ADR431 2.5V REFERENCE

Figure 62. Low-Z 2.5 V Buffer

Other Impact of Mismatches in the Feedback Networks
The internal common-mode feedback network still forces the output voltages to remain balanced, even when the RF/RG feedback networks are mismatched. However, the mismatch will cause a gain error proportional to the feedback network mismatch. Ratio-matching errors in the external resistors degrade the ability to reject common-mode signals at the VAN and VIN input terminals, much the same as with a four-resistor difference amplifier made from a conventional op amp. Ratio-matching errors also produce a differential output component that is equal to the VOCM input voltage times the difference between the feedback factors (汕s). In most applications using 1% resistors, this component amounts to a differential dc offset at the output that is small enough to be ignored.

Another way to avoid the input common-mode swing limitation is to use dual power supplies on the AD8139. In this case, the biasing circuitry is not required.

Bandwidth vs. Closed-Loop Gain
The 3 dB bandwidth of the AD8139 decreases proportionally to increasing closed-loop gain in the same way as a traditional voltage feedback operational amplifier. For closed-loop gains greater than 4, the bandwidth obtained for a specific gain can be estimated as

f ? 3 dB,VOUT , dm =

RG ℅ (300 MHz) RG + RF

(20)

or equivalently, 汕(300 MHz). This estimate assumes a minimum 90∼ phase margin for the amplifier loop, which is a condition approached for gains greater than 4. Lower gains show more bandwidth than predicted by the equation due to the peaking produced by the lower phase margin.

Rev. B | Page 21 of 24

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AD8139
Driving a Capacitive Load
A purely capacitive load reacts with the bondwire and pin inductance of the AD8139, resulting in high frequency ringing in the transient response and loss of phase margin. One way to minimize this effect is to place a small resistor in series with each output to buffer the load capacitance (see Figure 58 and Figure 63). The resistor and load capacitance form a first-order, low-pass filter; therefore, the resistor value should be as small as possible. In some cases, the ADCs require small series resistors to be added on their inputs.
5 RS = 30.1? 4 CL = 15pF 3 2 1 0 每1 每2 RS = 60.4? 每3 CL = 15pF 每4 每5 每6 每7 RS = 60.4? 每8 CL = 5pF 每9 VS = ㊣5V 每10 V = 0.1V p-p 每11 GO, dm = 1 (RF = RG = 200?) 每12 R L, dm = 1k? 每13 10M 100M FREQUENCY (Hz) RS = 30.1? CL = 5pF

The input resistance presented by the AD8139 input circuitry is seen in parallel with the termination resistor, and its loading effect must be taken into account. The Thevenin equivalent circuit of the driver, its source resistance, and the termination resistance must all be included in the calculation as well. An exact solution to the problem requires the solution of several simultaneous algebraic equations and is beyond the scope of this data sheet. An iterative solution is also possible and simpler, especially considering the fact that standard 1% resistor values are generally used. Figure 64 shows the AD8139 in a unity-gain configuration driving the AD6645, which is a 14-bit, high speed ADC, and with the following discussion, provides a good example of how to provide a proper termination in a 50 次 environment. The termination resistor, RT, in parallel with the 268 次 input resistance of the AD8139 circuit (calculated using Equation 19), yields an overall input resistance of 50 次 that is seen by the signal source. To have matched feedback loops, each loop must have the same RG if they have the same RF. In the input (upper) loop, RG is equal to the 200 次 resistor in series with the (+) input plus the parallel combination of RT and the source resistance of 50 次. In the upper loop, RG is therefore equal to 228 次. The closest standard 1% value to 228 次 is 226 次 and is used for RG in the lower loop. Greater accuracy could be achieved by using two resistors in series to obtain a resistance closer to 228 次. Things get more complicated when it comes to determining the feedback resistor values. The amplitude of the signal source generator VS is two times the amplitude of its output signal when terminated in 50 次. Therefore, a 2 V p-p terminated amplitude is produced by a 4 V p-p amplitude from VS. The Thevenin equivalent circuit of the signal source and RT must be used when calculating the closed-loop gain, because in the upper loop, RG is split between the 200 次 resistor and the Thevenin resistance looking back toward the source. The Thevenin voltage of the signal source is greater than the signal source output voltage when terminated in 50 次 because RT must always be greater than 50 次. In this case, RT is 61.9 次 and the Thevenin voltage and resistance are 2.2 V p-p and 28 次, respectively. Now the upper input branch can be viewed as a 2.2 V p-p source in series with 228 次. Because this is a unitygain application, a 2 V p-p differential output is required, and RF must therefore be 228 ℅ (2/2.2) = 206 次. The closest standard value to this is 205 次. When generating the Typical Performance Characteristics data, the measurements were calibrated to take the effects of the terminations on the closed-loop gain into account.

CLOSED LOOP GAIN (dB)

RS = 0? CL, dm = 0pF

1G

Figure 63. Frequency Response for Various Capacitive Load and Series Resistance

The Typical Performance Characteristics that illustrate transient response vs. the capacitive load were generated using series resistors in each output and a differential capacitive load.

Layout Considerations
Standard high speed PCB layout practices should be adhered to when designing with the AD8139. A solid ground plane is recommended, and good wideband power supply decoupling networks should be placed as close as possible to the supply pins. To minimize stray capacitance at the summing nodes, the copper in all layers under all traces and pads that connect to the summing nodes should be removed. Small amounts of stray summing-node capacitance cause peaking in the frequency response, and large amounts can cause instability. If some stray summing-node capacitance is unavoidable, its effects can be compensated for by placing small capacitors across the feedback resistors.

Terminating a Single-Ended Input
Controlled impedance interconnections are used in most high speed signal applications, and they require at least one line termination. In analog applications, a matched resistive termination is generally placed at the load end of the line. This section deals with how to properly terminate a single-ended input to the AD8139.

04679-076

Rev. B | Page 22 of 24

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AD8139
Because this is a single-ended-to-differential application on a single supply, the input common-mode voltage swing must be checked. From Figure 64, 汕 = 0.52, VOCM = 2.4 V, and VICM is 1.1 V p-p swinging about ground. Using Equation 16, VACM is calculated to be 0.53 V p-p swinging about a baseline of 1.25 V, and the minimum negative excursion is approximately 1 V.

Exposed Paddle (EP)
The 8-lead SOIC and the 8-lead LFCSP have an exposed paddle on the bottom of the package. To achieve the specified thermal resistance, the exposed paddle must be soldered to one of the PCB planes. The exposed paddle mounting pad should contain several thermal vias within it to ensure a low thermal path to the plane.

5V

3.3V

0.01?F 0.01?F 205? 50? VS 2V p-p RT 61.9? 200? VOCM 3 8 2 1 226? + 5 25? AIN AVCC DVCC

0.01?F

SIGNAL SOURCE

AD8139
每 6 4 AIN 205? 25? GND C1

AD6645

C2 0.1?F 0.1?F

VREF
04679-054

2.4V

Figure 64. AD8139 Driving AD6645, 14-Bit, 80 MSPS/105 MSPS ADC

Rev. B | Page 23 of 24

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AD8139 OUTLINE DIMENSIONS
4.00 (0.157) 3.90 (0.154) 3.80 (0.150) 5.00 (0.197) 4.90 (0.193) 4.80 (0.189)
8 1 5 4

2.29 (0.090)

TOP VIEW

6.20 (0.244) 6.00 (0.236) 5.80 (0.228) BOTTOM VIEW
(PINS UP)

2.29 (0.090)

1.27 (0.05) BSC 1.75 (0.069) 1.35 (0.053) 0.10 (0.004) MAX COPLANARITY 0.10 1.65 (0.065) 1.25 (0.049) SEATING PLANE

0.50 (0.020) 0.25 (0.010)

45∼

0.51 (0.020) 0.31 (0.012)

0.25 (0.0098) 0.17 (0.0067)

8∼ 0∼

1.27 (0.050) 0.40 (0.016)

COMPLIANT TO JEDEC STANDARDS MS-012-A A
060506A

CONTROLLING DIMENSIONS ARE IN MILLIMETER; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.

Figure 65. 8-Lead Standard Small Outline Package with Exposed Pad [SOIC_N_EP] Narrow Body (RD-8-1)〞Dimensions shown in millimeters and (inches)
3.25 3.00 SQ 2.75 0.60 MAX 0.60 MAX
5 8

0.50 BSC

PIN 1 INDICATOR

TOP VIEW

2.95 2.75 SQ 2.55

EXPOSED PAD
(BOTTOM VIEW)

1.60 1.45 1.30 PIN 1 INDICATOR

4

1

12∼ MAX 0.90 MAX 0.85 NOM SEATING PLANE

0.70 MAX 0.65 TYP

0.50 0.40 0.30 0.05 MAX 0.01 NOM

1.89 1.74 1.59

Figure 66. 8-Lead Lead Frame Chip Scale Package [LFCSP_VD] 3 mm ℅ 3 mm Body, Very Thin, Dual Lead (CP-8-2)〞Dimensions shown in millimeters

ORDERING GUIDE
Model AD8139ARD AD8139ARD-REEL AD8139ARD-REEL7 AD8139ARDZ 1 AD8139ARDZ-REEL1 AD8139ARDZ-REEL71 AD8139ACP-R2 AD8139ACP-REEL AD8139ACP-REEL7 AD8139ACPZ-R21 AD8139ACPZ-REEL1 AD8139ACPZ-REEL71
1

061507-B

0.30 0.23 0.18

0.20 REF

Temperature Range 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C 每40∼C to +125∼C

Package Description 8-Lead Small Outline Package with Exposed Pad (SOIC_N_EP) 8-Lead Small Outline Package with Exposed Pad (SOIC_N_EP) 8-Lead Small Outline Package with Exposed Pad (SOIC_N_EP) 8-Lead Small Outline Package with Exposed Pad (SOIC_N_EP) 8-Lead Small Outline Package with Exposed Pad (SOIC_N_EP) 8-Lead Small Outline Package with Exposed Pad (SOIC_N_EP) 8-Lead Lead Frame Chip Scale Package (LFCSP_VD) 8-Lead Lead Frame Chip Scale Package (LFCSP_VD) 8-Lead Lead Frame Chip Scale Package (LFCSP_VD) 8-Lead Lead Frame Chip Scale Package (LFCSP_VD) 8-Lead Lead Frame Chip Scale Package (LFCSP_VD) 8-Lead Lead Frame Chip Scale Package (LFCSP_VD)

Package Option RD-8-1 RD-8-1 RD-8-1 RD-8-1 RD-8-1 RD-8-1 CP-8-2 CP-8-2 CP-8-2 CP-8-2 CP-8-2 CP-8-2

Branding

HEB HEB HEB HEB# HEB# HEB#

Z = RoHS Compliant Part, # denotes RoHS product may be top or bottom marked.

?2007 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D04679-0-10/07(B)

Rev. B | Page 24 of 24


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